Charge compensation control circuit and method for use with output driver

ABSTRACT

An output driver has an output multiplexor and an output current driver. The output multiplexor receives a data signal and outputs a q-node signal. The output current driver receives the q-node signal and drives a bus based on the q-node signal. The output multiplexor processes the data signal in various ways to generate the q-node signal. The output current driver is responsive to current control bits to select a amount of output drive current. In addition, the output multiplexor is controlled such that the output impedance of the output current driver is maintained within a predetermined range.

This application is a continuation of U.S. patent application Ser. No.09/222,590, filed Dec. 28, 1998, now U.S. Pat. No. 6,163,178.

BACKGROUND OF THE INVENTION

The present invention relates generally to an output driver forintegrated circuits, and more specifically to an apparatus and methodfor a bus output driver for integrated circuits.

Integrated circuits connect to and communicate with each other.Typically, integrated circuits communicate with each other using a buswith address, data and control signals.

In FIG. 1, a bus 20 interconnects a memory controller 22 and memorymodules (RAMS) 24, 26, 28. Physically, the bus comprises the traces on aprinted circuit board, wires or cables and connectors. Each of theseintegrated circuits has a bus output driver circuit 30 that interfaceswith the bus 20 to drive data signals onto the bus to send data to otherones of the integrated circuits. In particular, the bus output drivers30 in the memory controller 22 and RAMS 24, 26, 28 are used to transmitdata over the bus 20. The bus 20 operates or transmits signals at aspeed that is a function of many factors such as the system clock speed,the bus length, the amount of current that the output drivers can drive,the supply voltages, the spacing and width of the wires or traces makingup the bus, the physical layout of the bus itself and the resistance ofa pull up resistor attached to each bus.

The address, data and control lines making up the bus will be referredto as channels. In some systems, all channels connect to a pull-upresistor Z₀. Typically the resistance of the pull up resistor is 28ohms.

Output drivers for use on a bus, such as is shown in FIG. 1, arepreferably current mode drivers, which are designed to drive the bus 20with a determinable amount of current substantially independent of thevoltage on the driver output. The output impedance of the driver 30 is agood metric of how much the driver's current will change with voltagechanges on the driver's output, a high output impedance being desirablefor the current mode driver. In addition, a high output impedance isdesirable to minimize transmission line reflections on the bus when aparticular driver 30 receives voltage changes from another driver on thebus 20. In such a case, a driver with a high output impedance will notsubstantially alter the impedance of the bus 20, thus causing only asmall portion of a wave to be reflected at the location where the driver30 is attached to the bus.

FIG. 2A shows a prior art bus output driver circuit 30 which has anoutput multiplexor 32 that connects to an output current driver 34 atq-node 40. The q-node 40 refers to the physical connection between theoutput multiplexor 32 and the output current driver 34. A q-node signalis output to the q-node 40. The q-node signal is a voltage level thatcauses the output current driver 34 to drive a corresponding voltagelevel on the bus 20 (also herein called a bus channel).

The output multiplexor 32 receives a clock signal at a clock input 42,and receives odd and even data signals at the odd data and even datainputs 44 and 46, respectively. The odd and even data signals aresynchronized to the clock signal. The output multiplexor 32 transmitsthe data from the odd data and even data inputs onto the q-node 40 onthe rising and falling edges of the clock signal, respectively.

The slew rate and output current of the bus output driver 30 arecontrollable. A set of slew rate control bits 50 is used to select theslew rate of the transitions of the q-node signal. A slew rate estimator48 may be used to generate the slew rate control bits 50. Alternately,the slew rate control bits 50 may be generated by a process detector, aregister that is programmed with a fixed value during manufacture orduring testing of the device after manufacture, or by any other type ofslew rate detection circuitry. The source of the slew rate control bits50 may be external to the bus output driver 30. The output currentdriver 34 outputs a signal, called Vout, that corresponds to the q-nodesignal, onto the bus channel 20. A current control block 52 outputs aset of current control bits 54 that select the amount of current used todrive data onto the bus channel 20. The current control block 52 may beexternal to the bus output driver 30, and may be implemented as acurrent level detector or as a register programmed with a fixed valueduring or after manufacture of the device.

FIG. 2B is a schematic of the prior art output multiplexor and outputcurrent driver of FIG. 2A. The clock, odd data and even data signals areinput to multiple current control blocks 62, 64 and each current controlblock 62, 64 outputs a q-node signal on a q-node 66, 68, 70, 72. In FIG.2B, the q-nodes 66, 68, 70, 72 are also designated as q<6>−q<0>,respectively. When the q-node signal has a sufficiently high voltagelevel, a corresponding transistor T₀-T₆ in the output current driverbecomes active and pulls Vout low. Each q-node signal drives a binaryweighted pulldown device T₀-T₆ in the output current driver. In otherwords, multiple q-nodes 66, 68, 70, 72 are used to drive a singlechannel 20 of the bus. The transistors T₀-T₆ are n-typemetal-oxide-semiconductor (MOS) transistors and are binary weighted withrespect to each other. In particular, each transistor T₀-T₆ will driveor sink a predetermined amount of current with respect to I_(out).Transistor T₀ sinks 2⁰ or 1×I_(out) (e.g., about 0.26 milliampsminimum), transistor T₄ sinks 2⁴ or 16×I_(out), transistor T₅ sinks32×I_(out), and transistor T₆ sinks 64×I_(out).

Since the current control blocks 62, 64 are similar, one current controlblock 62 will be described. The current control block 62 has an inputblock 82 and a pre-driver 84. The input block 82 is responsive to acurrent control signal output on a current control bit line 84. In FIG.2B, the current control signals are shown as Current Control <0> throughCurrent Control <6>. Each q-node 66, 68, 70, 72 is associated with aseparate current control signal. The current control signal enables theNAND gates 86, 88 to respond to the odd and even data signals. Each NANDgate 86, 88 outputs, its signal to a pair of passgates 92, 94,respectively. The passgate pairs 92, 24 are responsive to the clocksignal such that one passgate pair 92, 94 is on at a time, outputtingeither the odd data or even data signal. The output of the passgates 92,94 is connected to the pre-driver 84.

If the current control signal on the current control bit line 84 is at alow voltage, the NAND gates 86, 88 output a high voltage levelregardless of the voltage level of the odd or even data signal, therebycausing a “low” voltage level at the associated q-node and disabling thecorresponding transistor in the output current driver.

If the current control signal on the current control bit line 84 is at ahigh voltage level, the NAND gates 86, 88 are enabled, and the predriver84, q-node and output current driver are responsive to the odd and evendata signals.

In the prior art output driver 30, the output impedance of the outputdriver 30 is not well controlled, and is determined by the value of asupply voltage, Vcc (the high voltage for the q-node), the outputvoltage when it is being driven low, and the characteristics of thetransistors in the output current driver 34.

FIG. 2C is a schematic of the prior art pre-driver 84 of FIG. 2B. Thepredriver has many predriver sub-blocks 96, 98, 100. Each predriversub-block 96, 98, 100 has an inverter I1, I2, I3 and a passgate pair102, 104, 106 respectively. One predriver sub-block 96 is always enabledwith the gate of each transistor of the passgate pair 102 connected tothe power supply Vcc and to ground, respectively. The other passgatepairs 104, 106 of the predriver sub-blocks 98, 100 connect to the slewrate control bits, Slew Rate Control <0> and Slew Rate Control <1>. Theslew rate of the predriver 84 is adjusted by enabling and disabling thepassgates 104, 106 with slew rate control signals on the slew ratecontrol bits.

In particular, when the slew rate control signal on Slew Rate Controlbit <1> is high, the passgate pair 104 of the predriver subblock 98 isenabled. The passgate pair 104 increases the rate of transition betweena high voltage level and a low voltage level of the q-node signal on theq-node 66. When the slew rate control bit <1> is low, the correspondingpassgate pair 104 of the predriver sub-block 98 is effectively disabledand the slew rate is unaffected. Enabling the additional passgate pairsof additional predriver sub-blocks 100 further increases the slew rateof the q-node signal.

However, when using multiple q-nodes to drive a single channel, it isdifficult to match the delays and slew rates of each q-node under allprocess, voltage and temperature conditions.

Therefore, there is a need for an output driver whose output impedanceis maintained within a desired range. There is also a need for animpedance controlled output driver which ha s an adjustable slow rateand operating current.

SUMMARY OF THE INVENTION

An output driver has an output multiplexor and an output current driver.The output multiplexor receives a data signal and outputs a q-nodesignal to the output current driver. In the output current driver, anoutput drive transistor receives the q-node signal. The output drivetransistor has a predetermined threshold voltage and an output impedancewhich is maintained within a predetermined range when the output drivetransistor is outputting a low voltage level.

In this way, the q-node signal is used to control the slew rate andoutput impedance of the signal output by the output driver.

In another embodiment, the output current driver is responsive to acurrent control signal which is used to select a desired amount of drivecurrent onto the bus. The output current driver has transistor stacksthat are responsive to the current control signal to enable the q-nodesignal to cause a predetermined amount of current to flow through thetransistor stack.

From another viewpoint, the present invention is directed to a methodand apparatus that satisfies the need to have an output driver with anadjustable operating current and adjustable slew rate.

In a preferred embodiment, the output driver includes an output currentdriver operating as a current mode driver. The output current driver isdriven from a predriver which receives its power from a carefullyregulated power supply. The regulated supply causes the high voltagelevel of the control signal to be substantially equal to the regulatedsupply voltage in order to maintain the output impedance of the outputcurrent driver above a predetermined threshold when the output driver isoutputting a low voltage level. Additionally, the output current driverincludes circuitry to permit the operating current of the output currentdriver to be adjustable and the predriver includes circuitry to permitthe slew rate of the control signal to be adjustable. To help meet thegoals of an adjustable slew rate and adjustable operating current, asingle control node (q-node) is employed between the predriver and theoutput current driver. Not only does the single control node simplifyimplementation of the adjustable operating current and adjustable slewrate features, it further simplifies the design of the impedancecontrolled driver. Thus, the output driver has a controlled anddeterminable output impedance. Additionally, an impedance controlleddriver has an adjustable slew rate and adjustable operating current. Theresult is a driver having controlled switching characteristics, a morestable output current on a bus, and a driver which minimizes reflectionsfrom other drivers on the bus. The regulated power supply for thepredriver includes a v-gate supply for generating the regulated supplyvoltage. A charge compensator is coupled to the predriver to helpmaintain the v-gate supply voltage when the predriver is changing state.The v-gate supply includes a v-gate generator for generating theregulated supply voltage and a charge compensation bit generator forcontrolling the charge compensator. Furthermore, to maintain the dutycycle of the output signal from the output driver when slew rateadjustments are made, a duty cycle compensator is employed topre-compensate the signal received by the predriver. Also, to aid thepredriver in driving the control signal to ground, a kickdown circuit isemployed and coupled in parallel with the predriver.

BRIEF DESCRIPTION OF THE DRAWINGS

Additional objects and features of the invention will be more readilyapparent from the following detailed description and appended claimswhen taken in conjunction with the drawings, in which:

FIG. 1 is a block diagram of a prior art bus connecting a memorycontroller and RAMS.

FIG. 2A is a block diagram of a prior art bus output driver with anoutput multiplexor and output current driver.

FIG. 2B is a schematic of the prior art output multiplexor and outputcurrent driver of FIG. 2A.

FIG. 2C is a schematic of a prior art pre-driver of FIG. 2B.

FIG. 3 is a block diagram of a bus output driver of the presentinvention.

FIG. 4 is a detailed block diagram of the bus output driver of FIG. 3.

FIG. 5 is a schematic of the output current driver of FIG. 4.

FIG. 6 is a schematic of the pre-driver of FIG. 4.

FIG. 7A is a schematic of the kickdown circuit of FIG. 4.

FIG. 7B depicts the waveform of the q-node signal during a transitionfrom a high voltage level, Vcc, to the low voltage level, ground, usingthe kickdown circuit of FIG. 7A.

FIG. 8 is a schematic of the duty cycle compensator of FIG. 4.

FIG. 9 is a schematic of a preferred embodiment of the pre-driver ofFIG. 4.

FIG. 10A is a diagram of the charge compensator of FIG. 4.

FIG. 10B is a schematic of an alternate embodiment of the chargecompensator of FIG. 10A and uses a fixed width pulse to adjust thecharge on the V-gate supply.

FIG. 10C is a schematic of one embodiment of the rising edge detector ofFIG. 10A.

FIG. 10D is a schematic of an alternate embodiment of the rising edgedetector of FIG. 10A.

FIG. 10E is a schematic of an alternate embodiment of the rising edgedetector and charge compensation bit generator of FIG. 10A and FIG. 4,respectively.

FIG. 10F is a schematic of the tri-state inverters of FIG. 10E.

FIG. 10G is a schematic of another alternate embodiment of the risingedge detector of FIG. 10A.

FIG. 10H is a schematic of the tri-state inverters of FIG. 10G.

FIG. 11A is a schematic of the V-gate generator of FIG. 4.

FIG. 11B is a model circuit used to determine the value of V-gate thatis output by the V-gate generator of FIG. 11A.

FIG. 11C is a schematic of a V-gate reference voltage generator of FIG.11A.

FIG. 12A is a block diagram of the charge compensation bit generator ofFIG. 4, and FIG. 12B is a state diagram for a finite state machine inthe charge compensation bit generator.

FIG. 13 is a schematic of the regulator of the charge compensation bitgenerator of FIG. 12A.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

In FIG. 3, a bus output driver 120 has an output multiplexor 122 thatconnects to an output current driver 124 at a q-node 126. A clocksignal, an even data signal, and an odd data signal are supplied to theoutput multiplexor 122 at a clock input 127, an even data input 128 andan odd data input 129, respectively. The output multiplexor 122 outputsa q-node signal on the q-node 126 that is used to control the slew rateand output impedance of a channel signal, Vout. In response to theq-node signal, the output current driver 124 outputs the channel signal,Vout, on the channel 130 of the bus. A V-gate supply block 132 suppliescontrol signals and a V-gate voltage to the output multiplexor 122 andwill be discussed in detail below. In another embodiment, the V-gatesupply block 132 is not used.

The output multiplexor 122 is responsive to a slew rate control signal,consisting of slew rate control (SRC) bits 135, received from a slewrate estimator, which may be external to the bus output driver 120. Theslew rate estimator is not part of the present invention, but is part ofthe context in which the invention operates.

The output current driver 124 is responsive to current control bits 136from a current control block, which may be external to the bus outputdriver 120. The current control block is not part of the presentinvention, but is part of the context in which the invention operates.

FIG. 4 shows the overall architecture of the bus output driver 120 ofthe present invention. In particular, the output multiplexor 122 has thefollowing components:

An input block 140, with dual pass-gate pairs 142, 144, receives andmultiplexes the data signals using the clock signal. The passgate pairsmultiplex the odd and even data signals using the clock signal to outputa clocked data signal.

A duty cycle compensator 146 generates a precompensated clocked datasignal by modifying the duty cycle of the clocked data signal by apredetermined amount.

A predriver 148 generates a q-node signal by selectively modifying theslew rate of the precompensated clocked data signal. The q-node signaltransitions between a low voltage level and a high voltage level and hasa duty cycle which results in a 50% duty cycle at Vout. The predriver isresponsive to slew rate control signals on the slew rate control bits135.

A kickdown circuit 150 increases the rate of transition of the q-nodesignal from a high voltage level to a low voltage level for a portion ofthe transition.

A charge compensator 152 maintains a V-gate voltage and hence the highvoltage level of the q-node signal within a predetermined range. Thecharge compensator 152 delivers a predetermined amount of charge to theV-gate voltage based on a rising or falling edge transition of theincoming data signal to the predriver.

The V-gate supply block 132 has a charge compensation bit generator 156and a V-gate generator 158. A single V-gate supply block 132 suppliesthe V-gate voltage and charge compensation bits for multiple outputmultiplexors 122. In an alternate embodiment, the V-gate supply block132 is a part of each output multiplexor 122.

The V-gate generator 158 supplies the V-gate voltage to the predriver148. The V-gate voltage is different from a supply voltage Vcc.Consequently, the high voltage level of the q-node signal output by thepredriver 148 is substantially equal to the V-gate voltage.

The charge compensation bit generator 156 generates charge compensationcontrol signals, also called charge compensation bits, to control theamount of charge delivered by the charge compensator 152 to the V-gatevoltage supply.

In the circuit diagrams in this document, the triangular circuit groundsymbol is used to indicate the circuit Vss node, and the horizontal barsymbol for the power supply is used to indicate the circuit Vcc node,unless otherwise indicated.

Output Current Driver

FIG. 5 is a schematic of the output current driver 124 of FIG. 4. Theoutput current driver 124 includes multiple transistor stacks 162-174connected in parallel between the channel 130 and ground. Eachtransistor stack 162-174 has two n-type transistors, an upper transistorT₁₀, T₁₂, T₁₄, T₁₆, T₁₈, T₂₀ and T₂₂ and a lower transistor T₁₁, T₁₃,T₁₅, T₁₇, T₂₁, and T₂₃, respectively, that are connected in series. Theq-node signal is input to the gate of the upper or output drivetransistors T₁₀, T₁₂, T₁₄, T₁₆, T₁₈, T₂₀ and T₂₂. Current controlsignals on a set of current control bits, CC<0> through CC<6>, are inputto the gate of the lower transistor T₁₁, T₁₃, T₁₅, T₁₇, T₂₁ and T₂₃.When each of the current control signals is at or exceeds the thresholdvoltage (Vth) of the lower transistor, the corresponding lowertransistor T₁₁, T₁₃, T₁₅, T₁₇, T₂₁ and T₂₃ is enabled or “on.” When thelower transistor T₁₁, T₁₃, T₁₅, T₁₇, T₂₁ and T₂₃ is enabled and when theq-node signal transitions high (i.e., to its logic high voltage), apredetermined amount of current flows through the transistor stack162-174 to the circuit ground. Therefore, the output drive current isadjusted by setting a subset of the current control signals to a highvoltage level. Preferably, the lower transistors have a low, positivethreshold voltage Vt of about 0.3 volts, and more generally in the rangeof 0.3 to 0.4 volts. Alternately, the lower transistors have a normalthreshold voltage Vt of 0.7 volts or ranging from 0.6 to 0.8 volts.

To further provide a programmable output drive current, at least one ofthe transistor stacks 162-174 is binary weighted with respect to atleast one other transistor stack 162-174. Preferably the transistorpairs in all the transistor stacks of the output current driver 124 aresized so that the current drive capability of the transistor stacks 162,164, 166, 168, 170, 172 have current drive ratios of 64:32:16:8:4:2:1,respectively (i.e., are binary weighted). The transistors in the outputcurrent driver of the present invention can have a reduced channellength with respect to the prior art output driver of FIG. 2B.

Pre-Driver with Adjustable Slew Rate

In FIG. 6, the predriver 148 includes a base block 202 with at least oneslew rate adjustment block 204, 206. The base block 202 is alwaysenabled and outputs a q-node signal that has an associated,predetermined slew rate. The base block 202 has two invertors I4, I5connected in series which are sized to provide both an appropriate slewrate and duty cycle.

At least one controllable slew rate adjustment block 204, 206 isconnected in parallel with the base block 202. In one embodiment, theslew rate adjustment blocks 204, 206 each use the same circuit design.The slew rate adjustment blocks 204, 206 have a control block 212connected in series with a stacked transistor pair 214. The stackedtransistor pair 214 has a p-type MOS (p-type) transistor T₂₄ connectedin series with an n-type transistor T₂₅. The outputs 216, 218 of thestacked transistor pairs connect to the q-node 126. The control blocks212 are responsive to slew rate control signals, SRC<0> and SRC <1>,which enable the stacked transistor pair 214 to be responsive to thedata signal from the duty cycle compensator. The control blocks 212include a NAND gate 220 and a NOR gate 222. The NAND gate 220 enablesthe p-type transistor T₂₄ of the transistor stack 214 and the NOR gate222 enables the n-type transistor T₂₅ of the transistor stack 214.

If both slew rate adjustment blocks 204, 206 have their slew ratecontrol signals set to a high voltage level, the slew rate of the q-nodesignal at the q-node 126 is greater than if only one slew rateadjustment block has its slew rate control bit set.

In particular, when slew rate control bit zero SRC<0>is at a highvoltage level, the NAND gate 220 is enabled to be responsive to the datasignal from the duty cycle compensator, allowing the data signal todrive the upper p-type transistor T₂₄ of the transistor stack 214. Atthe same time, when SRC<0> is at a high voltage level, /SRC<0> is at alow voltage level which enables the NOR gate 222 to be responsive to thedata signal, allowing the data signal to drive the lower n-typetransistor T₂₅ of the transistor stack 214.

When the NAND and NOR gates, 220 and 222, respectively, are enabled, andwhen the data signal from the duty cycle compensator transitions to alow voltage level, a high voltage level appears at the output of the NORgate 222 that causes the lower n-type transistor T₂₅ to conduct currentto ground thereby increasing the rate at which the q-node 126 is drivento ground. At substantially the same time that a high voltage levelappears at the output of the NOR gate 222, a high voltage level appearsat the output of the NAND gate 220 that causes the upper p-typetransistor T₂₄ to not conduct current or “turn off.”

When the NAND and NOR gates, 220 and 222, respectively, are enabled, andwhen the data signal from the duty cycle compensator transitions to ahigh voltage level, a low voltage level appears at the output of theNAND 220 gate that causes the upper p-type transistor T₂₄ to conductcurrent thereby increasing the rate at which the q-node 126 is driven toa high voltage level. At substantially the same time as a low voltagelevel appears at the output of the NAND gate 220, a low voltage levelappears at the output of the NOR gate 222 that causes the lower n-typetransistor T₂₅ to turn off. When SRC<0> is at a low voltage level and/SRC<0> is at a high voltage level, the NAND and NOR gates, 220 and 222respectively, are not responsive to the data signal and are therebydisabled. Therefore, the transistor stack 214 is not responsive to theincoming data signal.

In one embodiment, one slew rate adjustment block 204 increases the slewrate by 0.5 with respect to the base block 202, while the other slewrate adjustment block increases the slew rate by 1.5 with respect to thebase block 202. However, the slew rate adjustment blocks 204, 206 canprovide other predetermined amounts of adjustment to the slew rate.

Comparing the pre-driver 148 of the present invention with the prior artcircuit of FIG. 2C, it is noted that the pre-driver 148 of the presentinvention has just one transistor between the pre-driver's power supplyand the q-node 126 output, while there are two transistors (one for thedriver inverter and one for the pass gate) in the prior art version. Asa result, drive transistors in the pre-driver 148 of the presentinvention can be smaller than those of the prior art for the same speedof operation, and their behavior in the linear region of operation canbe more easily controlled.

The slew rate adjustment blocks 204, 206 are sized to provide anappropriate slew rate without regard to the duty cycle to increase therange for each setting of the slew rate control bits. Therefore,activating the slew rate adjustment blocks will cause asymmetry in theduty cycle at in the output voltage Vout. The duty cycle compensator146, discussed below, compensates for this asymmetry.

Kickdown Circuit

FIG. 7A is a schematic of the kickdown circuit 150 of FIG. 4. Thekickdown circuit 150 aids the predriver 148 in driving the q-node signalto ground or Vss. The kickdown circuit 150 connects in parallel with thepredriver 148. The kickdown circuit 150 detects a falling edgetransition of the q-node signal on the q-node 126, then pulls the chargefrom the q-node 126 to ground (Vss) thereby increasing the rate at whichthe q-node 126 approaches Vss (ground), thereby assuring that the q-nodesignal swings fully to ground.

In the kickdown circuit 150, one transistor T₃₀ is responsive to theincoming data signal to the predriver 148 via inverter 16. The othertransistor T₃₁ is responsive to the q-node signal being output by thepredriver 148 via inverter 17.

The kickdown circuit 150 looks ahead to the incoming data to identifythe next data transition. When the q-node signal at the q-node 126 is ata high voltage level the lower transistor T₃₁ is off. When the incomingdata signal to the predriver 148 is at a low voltage level, the uppertransistor T₃₀ is on. As the q-node signal transitions to the incominglow voltage level, the lower transistor T₃₁ turns on and conductscurrent to ground.

FIG. 7B depicts the waveform of the q-node signal during a transitionfrom a high voltage level Vcc (or V-gate as will be seen with referenceto FIG. 9) to the low voltage level, ground. The solid line depicts theq-node signal using the kickdown circuit. The dotted line depicts theq-node signal without the kickdown circuit. Δt is the time differencebetween complete and incomplete voltage swings between Vcc and groundfor the q-node signal. Vtn is the threshold voltage for the nmos devicereceiving the q-node signal in the output current driver, and is equalto about 0.7 volts.

When the incoming data signal to the predriver is at a low voltage level(and thus turning on transistor T₃₀) and the q-node signal transitionsfrom a high voltage level past the “crossover” voltage V_(x) of inverter17, inverter 17 will output a sufficiently high voltage to turn ontransistor T₃₁. The cross-over voltage Vx of the inverter 17 depends onthe ratio of the transistors in the inverter, and is preferably set sothat transistor T₃₁ turns on when the q-node signal is about equal tothe transistor threshold voltage Vtn. In one embodiment, the crossovervoltage Vx is equal to about 0.5 times the supply voltage, Vcc.

When both transistors T₃₀, T₃₁ of the kickdown circuit are on andconducting current, the rate at which the q-node signal transitions tothe circuit ground voltage level is increased as shown in region 238.

When the incoming data signal to the predriver 148 is at a high voltagelevel, the kickdown circuit 150 is not enabled because transistor T₃₀ is“off”.

This technique obtains a full voltage swing between Vcc and ground onthe q-node signal without disturbing the slew rate of the q-node, andthus also Vout, as controlled by the predrivers. Full swings areimportant to guarantee constant delays through the output current driver124. When the voltage swings of the q-node signal are not full, datadependent delays occur and produce data dependent jitter of themagnitude of Δt shown in FIG. 7B.

Duty Cycle Compensator

FIG. 8 is a schematic of the duty cycle compensator 146 of FIG. 4. Theduty cycle compensator 146 pre-compensates for distortion of the dutycycle caused by the slew rate control blocks of the predriver 148 whenthe slew rate control (SRC) signals, SRC<1> and SRC<0>, are enabled. Inresponse to the slew rate control signals on the SRC bits, the dutycycle compensator 146 pre-compensates the data signals being input tothe predriver 148 such that the distortion caused by the predriver 148is canceled out in the q-node signal at the q-node 126. In other words,the duty cycle compensator 146 modifies the duty cycle of the clockeddata signal by a predetermined amount in response to the slew ratecontrol signals.

The duty cycle compensator 146 has a pair of series-connected invertorsI8, I9 and two transistor stacks 246, 248. The transistor stacks 246,248 have two n-type transistors T₃₂, T₃₃, T₃₄, T₃₅ connected in seriesbetween the input to the predriver 148 and ground. The input to theupper transistor T₃₂, T₃₄ is the signal output by the first inverter 18.The slew rate control bits connect to the gate of the lower transistorsT₃₃ and T₃₅ . A high voltage level on the slew rate control bits enablesthe stacked transistors 246, 248 to adjust the duty cycle of the clockeddata signal, by increasing the slew rate of high-to low transitions onthe input to the predriver 148. A low voltage level on the slew ratecontrol bits disables the stacked transistors 246, 248 and prevents theduty cycle of the clocked data signal from being modified.

To increase the range for each setting of the slew rate control bits,the transistors T₂₄ and T₂₅ (FIG. 6) of the slew rate control blocks aresized such that the rise and fall times of Vout are within a certainrange, regardless of the duty cycle. Therefore, activating the slew ratecontrol bits will cause asymmetry in the duty cycle of Vout. It isworthwhile to note that when Vout has a fifty percent duty cycle, theq-node signal does not have a fifty percent duty cycle.

Regulating the Q-node Voltage

FIG. 9 is a schematic of a preferred embodiment of the predriver 148 ofFIG. 4. The only difference between the predriver of FIG. 9 and FIG. 4is that the predriver of FIG. 9 is powered by a V-gate supply voltageinstead of the conventional power supply voltage Vcc. The V-gate supplyvoltage is a regulated voltage and is chosen such that the output drivetransistor of the output current driver 124 operates at the edge ofsaturation and causes the output impedance to exceed 150 ohms. In thisway, the q-node signal on the q-node 126 varies from a low level ofabout zero volts to a high level of about V-gate. In one preferredembodiment in which Vcc is equal to about 2.5 volts, V-gate is equal toabout 1.4 to 1.5 volts.

The output impedance exceeds 150 ohms for an range of supply voltagesfrom 2.25 volts to 2.75 volts, for a range of temperatures from about 0°C. to 110° C., and for a range of expected variation in transistorperformance variation.

When using the V-gate voltage to supply power to the predriver 148, theV-gate voltage may also, in some embodiments, be provided to a slew rateestimator (which may be external to the bus output driver) so that theslew rate estimator can more accurately predict the operation of thepredriver 148.

FIG. 10A is a diagram of the charge compensator 152 of FIG. 4. Thecharge compensator 152 reduces fluctuations in the V-gate voltage. Theamount of charge drawn by the predriver from the V-gate supply dependson whether the q-node signal is rising or falling and causes the V-gatevoltage to fluctuate. To reduce fluctuations in the V-gate voltage, thecharge compensator 152 has a rising edge detector 252 and a falling edgedetector 254. The detectors 252, 254 deliver a different amount ofcharge to the V-gate voltage supply line depending on whether theincoming data transition has a rising or a falling edge. The rising andfalling edge detectors 252 and 254 output a negative pulse, 256 and 258,respectively, when they detect their 30 respective edges. The negativepulses 256, 258 turn on p-type transistors T₃₆ and T₃₇ causing currentto flow from the supply voltage Vcc to the V-gate voltage supply linefor the duration of the pulse width 262, 264. T₃₆ and T₃₇ are sized toreflect the different amounts of charge compensation to be used for therising and falling edges, respectively. In a preferred embodiment, T₃₆is a 53 micron/0.25 micron p-channel transistor and T₃₇ is a 30micron/0.25 micron p-channel transistor, both having a threshold voltageof about 0.7 volts.

In a preferred embodiment, charge compensation signals on chargecompensation bits are also input to the rising and falling edgedetectors, 252 and 254, respectively. The charge compensation signalsselectively adjust the amount of charge output by the edge detectors252, 254.

The amount of charge from the charge compensator can be adjusted inseveral ways. For example, the circuit of FIG. 10B is similar to FIG.10A and uses a fixed width pulse to adjust the charge on the V-gatesupply. The charge compensation bits (CCB0 and CCB1) enable additionaltransistors T_(36A), T_(36B), T_(37A), T_(37B), that are connected inparallel with transistors T₃₆ and T₃₇, respectively, to provideadditional current to the V-gate supply when the corresponding chargecompensation bits, CCB0 and CCB1, are equal to “1”. In FIG. 10B, /CCB0and /CCB1 are the complement of CCB0 and CCB1 and are generated fromCCB0 and CCB1 using an inverter (not shown).

Other embodiments adjust the amount of charge by varying the length ofthe pulses. In a preferred embodiment of the rising edge detector, shownin FIG. 10C, the input data signal from the Duty Cycle Compensator (DCC)is supplied to a NAND gate 266 both directly and through an odd numberof invertors I10, I11, I12 to generate a pulse. The charge compensationbits control transistors T₃₈, T₃₉, T₄₀, T_(38A), T_(39A), T_(40A),T_(38B), T_(39B), T_(40B), to selectively add capacitors to increase theamount of delay of the input data signal through the inverter string andthereby increase the output pulse width from the NAND gate 266. In thisembodiment, a single charge compensation bit enables or disablesmultiple transistors, thus distributing the capacitive load and thedelay. A falling edge detector can be implemented in a similar manner.

Another embodiment of the rising edge detector is shown in FIG. 10D. Theembodiment shown in FIG. 10D is similar to the embodiment shown in FIG.10C. The input data signal from the duty cycle compensator is suppliedto an inverter I10.

In this embodiment, each charge compensation bit CCB0, CCB1, CCC2enables a single transistor-capacitor pair that is connected to theoutput of each inverter I10, I11 and I12, respectively. A falling edgedetector can be implemented in a similar manner.

FIG. 10E is another embodiment of the charge compensator of FIG. 10A. Inthis embodiment, the charge compensation bit generator 156 has digitalto analog (D/A) converters 268, 270 that convert the charge compensationbits to p-control and n-control signals, respectively. The control andn-control signals are analog signals. The inverters I10, I11 and I12 aretri-state inverters and are responsive to the p-control and n-controlsignals. Each tri-state inverter I10, I11 and I12 delays the signal byan amount in response to the p-control and n-control signals.

Referring also to FIG. 10F, a schematic of the inverters I10, I11 andI12 of FIG. 10E is shown. A pmos transistor T_(41A) and an nmostransistor T_(41B) are connected between the power supply Vcc and groundof an inverter I13. The p-control and n-control signals control theamount of current passed through transistors T_(41A) and T_(41B),respectively, thereby controlling the amount of delay of the output dataof inverter I13 with respect to the input data.

In FIG. 10G, yet another embodiment of the rising edge detector of FIG.10A is shown. Inverters I10, I11 and I12 connect in parallel to multipletri-state inverters. For example, inverter I10 connects in parallel toinverters I10A, I10B and I10C. Each charge compensation bit (CCB0, CCB1and CCB2) enables or disables a row of tri-state inverters. Note thatinverters I14, I15 and I16 generate the complement of CCB0, CCB1 andCCB2, as the en_p 0, en_p 1 and en_p 2 signals, respectively.

Referring also to FIG. 10H, a schematic of the tri-state inverters I10A,I11A, I12A, I10B, I11B, I12B, I10C, I11C and I12C of FIG. 10G is shown.The embodiment shown in FIG. 10H is similar to the embodiment shown inFIG. 10F. However, instead of having analog signals p-control andn-control, digital signals en_p and en_n control transistors T_(41A) andT_(41B), respectively, thereby turning the transistors T_(41A), T_(41B)on or off.

For example, CCB0 and its complement output by inverter I14 connect toinverters I10A, I11A and I12A via the en_n 0 and the en_p 0 inputs toenable or disable inverters I10A, I11A and I12A. When CCB0 is at a highvoltage level, en_n is at a high voltage level and en_p is at a lowvoltage level, thereby turning on transistors T_(41A) and T_(41B),respectively, and reducing the amount of delay and output pulse width.When CCB0 is at a low voltage level, en_n is at a low voltage level anden_p is at a high voltage level, thereby turning off transistors T_(41A)and T_(41B), respectively, and leaving the amount of delay and outputpulse width unchanged.

FIG. 11A is a schematic of the V-gate generator 158 of FIG. 4. TheV-gate generator 158 has a regulator 272, a capacitor 274 and a V-gatereference voltage generator 276. The regulator 272 is an operationalamplifier configured to match the V-gate voltage to the V-gate referencevoltage. The capacitor 274 reduces fluctuations in the V-gate supplyvoltage during periods of large current demands until the regulator 272responds.

The V-gate reference voltage generator 276 produces the V-gate referencevoltage that is input to the regulator 272. The V-gate reference voltageis chosen such that the output drive transistors of the output currentdriver remain on the edge of saturation when the output current driveris driving a signal with a low voltage level.

FIG. 11B shows an exemplary model circuit including the output drivetransistor T₁₀. Output drive transistor T₁₀ is also shown in FIG. 5, andthe following description assumes that the current control signal forthe output drive transistor T₁₀ is set such that the source of T₁₀ is atground. When outputting or driving a low voltage level V_(OL) to thechannel, the predriver drives the q-node signal at the gate of T₁₀ tothe V-gate voltage level. The drain of T₁₀ will then reach the V_(OL)voltage. To keep T₁₀ in saturation, V-gate minus the threshold voltage(Vth) of T₁₀ is less than or equal to the channel voltage, V_(OL). Thefollowing equation describes the relationship:

V-gate−Vth≦V_(OL),

which is equivalent to:

V-gate≦V_(OL)+Vth.

Since lower values of V-gate require larger dimensions for the outputdrive transistor, T₁₀, the maximum desirable voltage for V-gate issubstantially equal to V_(OL)+Vth. Operating the output drive transistorT₁₀ at the edge of saturation causes the output drive transistor T₁₀ tomaintain a high output impedance substantially equal to or exceedingabout 150 ohms, while keeping the size of the output drive transistorT₁₀ reasonably small.

In a preferred embodiment, V_(OL) is equal to about 0.8 volts, and theoutput transistor T₁₀ is an n-channel transistor having a thresholdvoltage of about 0.7 volts.

Referring back to FIG. 11A, an ideal V-gate reference generator 276 isshown. The V-gate reference generator has two voltage sources. Onevoltage source 278 generates a voltage substantially equal to V_(OL).The other voltage source 280 generates a voltage substantially equal tothe threshold voltage, Vth, of the output drive transistor, T₁₀.

FIG. 11C is a schematic of one embodiment of the V-gate referencegenerator 276 of the V-gate generator of FIG. 11A. In one embodiment,the lower voltage source 278 of FIG. 11A comprises a bandgap currentsource 282, a resistor 283 with resistance R, an operational amplifier284 and a transistor T₄₂. A current, I_(bandgap), from the bandgapreference current source 282 is dropped across resistor 283. The currentI_(bandgap) and the resistance R of the resistor 283 are chosen suchthat the resulting voltage V1 at node N1 substantially equals the lowlevel output voltage of the channel, V_(OL). Variations in theresistance R of the resistor 283 will be canceled by I_(bandgap) becausethe bandgap current source 282 is designed to change I_(bandgap) ininverse proportion to the resistance R of resistor 283, using techniqueswell known to those skilled in the art of designing bandgap references.The operational amplifier 284 or regulator along with n-type transistorT₄₂ drives the Vconst node to the voltage level V1 which substantiallyequals V_(OL).

Transistor T₄₄ represents the upper voltage source 280 of FIG. 11A.Another current source 286 generates a current called I_(bias) whichdoes not vary with resistance or temperature. The current I_(bias) flowsthrough transistor T₄₄. Transistor T₄₄ is an n-type transistor tied as adiode. Transistor T₄₄ is sufficiently large with respect to I_(bias)such the drain to sour e voltage across T₄₄ is close to the thresholdvoltage Vth of T₄₄.

Unfortunately, the currents, I_(bandgap) and I_(bias), vary withvoltage. This variance causes the V-gate reference voltage to beslightly higher than desired when the supply voltage Vcc is higher thanits nominal value, and lower than desired when the supply voltage Vcc islower than its nominal value.

To reduce this variation, in a preferred alternate embodiment, the lowervoltage source 278 of FIG. 11A also includes transistor T₄₆. TransistorT₄₆ is an n-type transistor with its gate tied to the supply voltageVcc. In this alternate embodiment, the voltage output as -from voltagesource 278 equals the voltage at node Vmid, which is substantially equalto V_(OL) plus the drain to source voltage (Vds) across T₄₆. The drainto source voltage Vds across T₄₆ will vary with Vcc. For a range of highsupply voltages, Vcc, the drain to source voltage Vds of T₄₆ willdecrease slightly. For a range of low supply voltages, Vcc the drain tosource voltage Vds of T₄₆ will increase slightly. Therefore, T₄₆ issized such that variations of Vds of T₄₆ and the variation of V1(R*I_(bandgap)) are canceled for a predefined range of supply voltages.In this way, a stable desired V-gate reference voltage is generated thatis substantially independent of voltage, process, resistivity andtemperature.

Referring to FIGS. 12A and 12B, the charge compensation bit generator156 outputs the appropriate charge compensation signals on the chargecompensation bits to the charge compensator. The charge compensation bitgenerator 156 has a regulator 292, a test output multiplexor 294 and afinite state machine (FSM) 296. Alternatively, t he charge compensationbits are driven by registers which can be accessed by other circuitry,or driven by other logic blocks or metal options.

The regulator 292 supplies a voltage called V-gate-test at node N2 tothe test output multiplexor 294. The regulator 292 is a scaled downversion of the V-gate generator which supplies the output multiplexorwith the V-gate voltage.

The test output multiplexor 294 duplicates many of the components ofFIG. 4 such as the input block 298, the duty cycle compensator 300,predriver 302, kickdown circuit 304, charge compensator 306 and testcharge compensation bits 308. The V-gate-test voltage is supplied to thepredriver 302 of the test output multiplexor 294. The even and odd datainputs of the input block 298 of the test output multiplexor 294 aretied to Vcc and ground respectively such that a q-node-test signaloutput from the predriver 302 toggles at every clock edge to create acontinuous current draw from the V-gate-test supply.

The test charge compensation signals on the test charge compensationbits 308 of the FSM 296 are set such that the net amount of currentbeing drawn from the regulator 292 that supplies V-gate-test at node N2will be equal to zero or as close to zero as possible. However, if thetest charge compensation signals are such that there isundercompensation, there will be a net current flow out of the regulator292. In contrast, if there is overcompensation, there will be a netcurrent flow into the regulator 292.

The regulator 292 adjusts for over or under compensation by providingcurrent to or drawing current from the V-gate-test voltage node N2. Theregulator 292 generates a “more comp signal”, that indicates whether theregulator 292 is providing or drawing current. The FSM 296 samples the“more comp signal” and updates the test charge compensation bits tocause the charge compensator 306 to change the amount of compensationaccordingly.

To determine a desired setting of the charge compensation signals, theFSM 296 iteratively changes the test charge compensation signals on thetest charge compensation bits. The test charge compensation bits arechanged n times, where n is the number of iterations needed to traversethrough all combinations of the charge compensation bits. The change inthe test charge compensation signals causes the amount of compensationprovided by the charge compensator 306 of the test output multiplexor294 to change, which in turn causes the regulator 292 to modify theV-gate-test voltage, and modifies the more comp signal. This procedurerepeats and at each iteration the regulator 292 provides lessmodification, until the test charge compensation bits are very close tooptimal.

The desired setting of the test charge compensation signals on the testcharge compensation bits 308 occurs when consecutive changes to thecharge compensation signals on the least significant bits of the testcharge compensation bits 308 causes the “more comp signal” to toggle. Atthis point, the value of the charge compensation bits of the outputmultiplexor 122 (FIG. 4) are updated with the same values as the testcharge compensation signals in the test output multiplexor 294.

An initiate locking signal is input to the FSM 296 on the initiatelocking input 310 to start the procedure to determine the desiredsetting of the charge compensation bits. The initiate locking signal isprovided at power on and at intervals as required to compensate forthermal voltage drift.

FIG. 13 is a schematic of the regulator 292 of the charge compensationbit generator of FIG. 12A. The regulator 292 is a two stage operationalamplifier. The operational amplifier attempts to keep the voltage atnode N2 equal to the voltage at node N3 which is the output of theV-gate reference block.

The operational amplifier connects to a comparator 312 which generatesthe “more comp signal.” Node A connects to the negative input of thecomparator 312 and node B connects to the positive input of thecomparator. If the voltage at V-gate-test, node N2, is below that of theV-gate reference voltage, node N3, then the voltage at node A fallsbelow that of node B, thereby turning on T₄₈ and providing more currentinto V-gate-test. If the voltage at V-gate-test, node N2, is above thatof the V-gate reference voltage, node N3, then the voltage at node Arises above that of node B, thereby turning off transistor T₄₈ andallowing transistor T₅₀ to draw current out of V-gate-test at node N2.

When the voltage at node A is less than the voltage at node B, morecompensation is needed from the charge compensators of the outputmultiplexor. When the voltage at node A exceeds the voltage at node B.less compensation is needed from the charge compensators of the outputmultiplexor. In this way, by comparing the voltages at nodes A and B,the comparator 312 outputs a more compensation signal having a firstvoltage level when more charge compensation is needed and a secondvoltage level when less charge compensation is needed.

While the present invention has been described with reference to a fewspecific embodiments, the description is illustrative of the inventionand is not to be construed as limiting the invention. Variousmodifications may occur to those skilled in the art without departingfrom the true spirit and scope of the invention as defined by theappended claims.

What is claimed is:
 1. A charge compensation control circuit,comprising: a first node operable to provide a supply voltage for afirst circuit having charge consumption characteristics; a first chargecompensation circuit operably coupled to the first node, wherein thefirst charge compensation circuit is operable to supply supplementalcharge to the first node based on a first charge compensation value; anda control circuit operably coupled to the first charge compensationcircuit, the control circuit including: a second node operable toprovide a supply voltage to a second circuit, wherein charge consumptioncharacteristics of the second circuit are substantially similar tocharge consumption characteristics of the first circuit; a second chargecompensation circuit operably coupled to the second node, wherein thesecond charge compensation circuit is operable to supply supplementalcharge to the second node based on a second charge compensation value; aregulator operably coupled to the second node, the regulator operativeto control a voltage at the second node such that the voltage at thesecond node corresponds to a reference voltage, wherein the regulatorgenerates a compensation signal when the regulator responds to changesin the voltage on the second node so as to maintain the second node atthe voltage corresponding to the reference voltage; and a control blockoperably coupled to the regulator and operable to generate the first andsecond charge compensation values, wherein the control block adjusts thefirst and second charge compensation values based on the compensationsignal.
 2. The charge compensation control circuit of claim 1, whereinthe second charge compensation value and the first charge compensationvalue are equal.
 3. The charge compensation control circuit of claim 1,wherein the control block is operable to adjust the second chargecompensation value such that modification of the voltage at the secondnode by the regulator is minimized.
 4. The charge compensation controlcircuit of claim 1, wherein the charge consumption characteristics ofthe first circuit are dependent on at least one data signal of the firstcircuit.
 5. The charge compensation control circuit of claim 4, whereinthe first circuit is a predriver in an output circuit and the at leastone data signal includes a data value to be output using the outputcircuit.
 6. The charge compensation control circuit of claim 1, whereinthe first charge compensation circuit is operable to supply thesupplemental charge to the first node by enabling a charge path for atime period corresponding to a pulse width, wherein the pulse width isbased on the first charge compensation value.
 7. The charge compensationcontrol circuit of claim 1, wherein the first charge compensationcircuit is operable to supply the supplemental charge to the first nodeby enabling a charge path having a variable flow rate for a time periodcorresponding to a pulse width, wherein the flow rate of the charge pathis based on the first charge compensation value.
 8. The chargecompensation control circuit of claim 1, wherein the control blockincludes a finite state machine.
 9. The charge compensation controlcircuit of claim 1, wherein the control block is operable to selectivelyupdate the first charge compensation value when the compensation signaltoggles.
 10. An apparatus for use in conjunction with an output driver,comprising: a predriver for generating a first signal based on a clockeddata signal; a voltage regulator for generating a regulated test voltageon a power supply node of the predriver for use as the high voltagelevel of the first signal, and for generating a second signal when thevoltage regulator adds charge to the power supply node to maintain thevoltage on the power supply node; a charge compensator for maintainingthe test voltage by supplying supplemental charge to the power supplynode; and a finite state machine for generating a test compensationsignal, wherein the charge compensator supplies an amount ofsupplemental charge to the power supply node based on the testcompensation signal, the finite state machine adjusting the testcompensation signal in response to the second signal and, uponcompletion of a predefined test sequence, outputting a compensationsignal corresponding to a final value of the test compensation signal.11. The apparatus or claim 10, further comprising: an input block formultiplexing an odd data signal and an even data signal with a clocksignal to output a clocked data signal; and a duty cycle compensator forreceiving the clocked data signal and outputting a precompensatedclocked data signal with a duty cycle that is changed by a predeterminedamount; wherein the predriver is configured to generate the first signalbased on the precompensated clocked data signal, the first signaltransitioning between a low voltage level and a high voltage level at apredetermined slew rate, such that the first signal has substantiallythe same duty cycle as the clocked data signal.